Researchers have discovered a way to capture and harness energy transmitted by such sources as radio and television transmitters, cell phone networks and satellite communications systems. By scavenging this ambient energy from the air around us, the technique could provide a new way to power networks of wireless sensors, microprocessors and communications chips.
"There is a large amount of electromagnetic energy all around us, but nobody has been able to tap into it," said Manos Tentzeris, a professor in the Georgia Tech School of Electrical and Computer Engineering who is leading the research. "We are using an ultra-wideband antenna that lets us exploit a variety of signals in different frequency ranges, giving us greatly increased power-gathering capability."
Tentzeris and his team are using inkjet printers to combine sensors, antennas and energy scavenging capabilities on paper or flexible polymers. The resulting self powered wireless sensors could be used for chemical, biological, heat and stress sensing for defense and industry; radio frequency identification (RFID) tagging for manufacturing and shipping, and monitoring tasks in many fields including communications and power usage.
A presentation on this energy scavenging technology was given July 6 at the IEEE Antennas and Propagation Symposium in Spokane, Wash. The discovery is based on research supported by multiple sponsors, including the National Science Foundation, the Federal Highway Administration and Japan's New Energy and Industrial Technology Development Organization (NEDO).
Communications devices transmit energy in many different frequency ranges, or bands. The team's scavenging devices can capture this energy, convert it from AC to DC, and then store it in capacitors and batteries. The scavenging technology can take advantage presently of frequencies from FM radio to radar, a range spanning 100 megahertz (MHz) to 15 gigahertz (GHz) or higher.
Scavenging experiments utilizing TV bands have already yielded power amounting to hundreds of microwatts, and multi-band systems are expected to generate one milliwatt or more. That amount of power is enough to operate many small electronic devices, including a variety of sensors and microprocessors.
And by combining energy scavenging technology with supercapacitors and cycled operation, the Georgia Tech team expects to power devices requiring above 50 milliwatts. In this approach, energy builds up in a battery-like supercapacitor and is utilized when the required power level is reached.
The researchers have already successfully operated a temperature sensor using electromagnetic energy captured from a television station that was half a kilometer distant. They are preparing another demonstration in which a microprocessor-based microcontroller would be activated simply by holding it in the air.
Exploiting a range of electromagnetic bands increases the dependability of energy scavenging devices, explained Tentzeris, who is also a faculty researcher in the Georgia Electronic Design Center at Georgia Tech. If one frequency range fades temporarily due to usage variations, the system can still exploit other frequencies.
The scavenging device could be used by itself or in tandem with other generating technologies. For example, scavenged energy could assist a solar element to charge a battery during the day. At night, when solar cells don't provide power, scavenged energy would continue to increase the battery charge or would prevent discharging.
Utilizing ambient electromagnetic energy could also provide a form of system backup. If a battery or a solar-collector/battery package failed completely, scavenged energy could allow the system to transmit a wireless distress signal while also potentially maintaining critical functionalities.
The researchers are utilizing inkjet technology to print these energy scavenging devices on paper or flexible paper-like polymers -- a technique they already using to produce sensors and antennas. The result would be paper-based wireless sensors that are self powered, low cost and able to function independently almost anywhere.
To print electrical components and circuits, the Georgia Tech researchers use a standard materials inkjet printer. However, they add what Tentzeris calls "a unique in house recipe" containing silver nanoparticles and/or other nanoparticles in an emulsion. This approach enables the team to print not only RF components and circuits, but also novel sensing devices based on such nanomaterials as carbon nanotubes.
When Tentzeris and his research group began inkjet printing of antennas in 2006, the paper-based circuits only functioned at frequencies of 100 or 200 MHz, recalled Rushi Vyas, a graduate student who is working with Tentzeris and graduate student Vasileios Lakafosis on several projects.
"We can now print circuits that are capable of functioning at up to 15 GHz -- 60 GHz if we print on a polymer," Vyas said. "So we have seen a frequency operation improvement of two orders of magnitude."
The researchers believe that self-powered, wireless
paper-based sensors will soon be widely available at very
low cost. The resulting proliferation of autonomous,
inexpensive sensors could be used for applications that
include:
Airport security: Airports have both multiple security
concerns and vast amounts of available ambient energy from
radar and communications sources. These dual factors make
them a natural environment for large numbers of wireless
sensors capable of detecting potential threats such as
explosives or smuggled nuclear material.
Energy savings: Self-powered wireless sensing devices
placed throughout a home could provide continuous
monitoring of temperature and humidity conditions, leading
to highly significant savings on heating and air
conditioning costs. And unlike many of today's sensing
devices, environmentally friendly paper-based sensors
would degrade quickly in landfills.
Structural integrity: Paper or polymer-based sensors
could be placed throughout various types of structures to
monitor stress. Self powered sensors on buildings, bridges
or aircraft could quietly watch for problems, perhaps for
many years, and then transmit a signal when they detected
an unusual condition.
Food and perishable material storage and quality
monitoring: Inexpensive sensors on foods could scan for
chemicals that indicate spoilage and send out an early
warning if they encountered problems.
Wearable bio-monitoring devices: This emerging wireless
technology could become widely used for autonomous
observation of patient medical issues.
USP 6917339
Multi-band broadband planar antennas
Multi-band broadband planar antennas
Inventor(s): LI RONGLIN [US]; TENTZERIS EMMANOUIL M [US]; LASKAR JOY [US] + (LI RONGLIN, ; TENTZERIS EMMANOUIL M, ; LASKAR JOY)
Classification: - international: H01Q1/24; H01Q11/12; H01Q21/30; H01Q5/00; H01Q7/00; H01Q9/30; H01Q9/42; (IPC1-7): H01Q11/12; H01Q7/00 - European: H01Q21/30; H01Q5/00B; H01Q9/30; H01Q9/42
Antennas of broadband and multi-band operation are presented. A broadband planar antenna includes two inverted-L antennas (ILAs) facing each other across a gap. One of the ILAs is input fed, and the other is electromagnetically coupled. The positioning of the gap affects the bandwidth. A dual-band planar antenna includes two ILAs facing each other across a gap with one of the ILAs being input fed, and the other being coupled. This dual-band planar antenna also includes a monopole antenna disposed between the two ILAs. A triple-band planar antenna includes two ILAs facing each other across a gap with one of the ILAs being input fed and the other IPA being coupled. This triple-band antenna also includes a monopole antenna disposed between the two ILAs, and a conductor extending horizontally from the monopole antenna towards, but not reaching the coupled ILA.; Another dual-band antenna includes an inner cut loop antenna encompassed by an outer cut loop antenna. Each of the cut loop antennas includes two ILAs with one of the ILAs being input fed and the other being coupled.
Description
RELATED APPLICATION
[0001] This application claims priority to and the benefit of the prior filed co-pending and commonly owned provisional patent application, which has been assigned U.S. Patent Application Ser. No. 60/413,327, entitled "Multi-band broadband planar wire antennas for wireless communication handheld terminals," filed on Sep. 25, 2002, and incorporated herein by this reference.
FIELD OF THE INVENTIONS
[0002] The inventions relate generally to antennas, and more particularly to planar antennas with multi-band and broadband functionalities such as may be used with mobile communication devices and in other compact antenna applications.
BACKGROUND OF THE INVENTIONS
[0003] In recent years, there has been a tremendous increase in the use of wireless communication devices. The increased use has filled or nearly filled existing frequency bands. As a result, new wireless frequency band standards are emerging throughout the world. For example, the existing 1<st > (1G) and 2<nd > (2G) generation cellular mobile communication systems operate at:
the AMPS (824-894 MHz) and PCS (1850-1990 MHz) bands in North America;
the GSM (880-960 MHz) and DCS (1710-1880 MHz) bands in Europe; and
the PDC (810-915 MHz) and PHS (1895-1918 MHz) bands in Japan. For future wireless communication systems, such as the emerging 3rd generation (3G) systems or beyond, new spectrum may be allocated around 2 GHz (e.g., already identified 1920-2170 MHz band for UMTS or IMT2000).
[0008] Like cellular mobile communications systems, Wireless Local Area Networks (WLANs) also use various frequency bands. IEEE 802.11b, Bluetooth, and HomeRF operate in the 2.4 GHz ISM band (2.400-2.485 GHz). IEEE802.11a and HiperLAN (in Europe) will use the 5 GHz ISM band (5.15-5.35 GHz and 5.725-5.825 GHz for IEEE802.11a, 5.15-5.25 GHz for HiperLAN1 and 5.15-5.35 GHz for HiperLAN2). Japan has started the development of standards for WLAN devices in the 5 GHz band.
[0009] As the frequency standards throughout the world change and evolve, wireless devices that can operate at the old and the new frequency standards are needed.
[0010] Increased functionality is another factor that drives the need for wireless devices that can operate at multiple frequencies. New wireless devices may provide multiple functions, but one or more of the functionalities may only be available at a respective one or more different frequencies from the base operating frequency. Thus, there is a need for wireless devices that can operate and implement functionalities at more than one frequency.
[0011] Yet another factor that drives the need for wireless devices that can operate at multiple frequencies is the desire of users for multi-functional services that operate at high data speeds including voice, video, and data transmissions. A wireless device may provide such services with automatic access and seamless roaming if the device can operate across multiple frequency bands.
[0012] The antenna is a key component in the realization of such a multi-mode wireless device. It is desirable for an antenna used in a multi-mode wireless device to include broadband performance for use in successive bands. It is also desirable for such an antenna to have multi-band performance for separated bands including far-separated bands. In addition to broadband and multi-band performance, it is desirable for such an antenna to be of a small size, a simple structure, and be of lightweight materials so as to be easily mounted in a handheld terminal with relatively low cost. Further, the radiation patterns in all service bands of such an antenna should be omni-directional and polarization-mixed to adapt to land-mobile propagation environments.
[0013] In recent years, a great number of new antenna structures have been developed for dual-band or triple-band operations in wireless communication handsets. A simple way to realize dual-band operation is to directly feed two antenna elements, each of which has a separate resonant frequency. For example, a combination of a monopole and a helical antenna, where the monopole is placed through the middle of the helix in the axial position and is simply connected to the end of the helix, has been successfully applied in GSM/DCS bands. Directly feeding two monopoles with different lengths can also result in two resonant frequencies. Another dual-band operation includes electromagnetically coupling two separate radiating elements. A coupling dual-band dipole antenna has been developed for WLAN applications in the 2.4 and 5.2 GHz bands. By coupling a rectangular element at the high frequency and an L-shaped element at the lower frequency, a dual-band operation was achieved for a planar inverted-F antenna (PIFA). The triple-band operation of the PIFA was implemented by adding one more L-shaped radiator.
[0014] Usually, a dual-band or triple-band antenna has a narrow bandwidth at each band. In order to achieve a broadband multi-band operation, some specific techniques or additional structures have to be incorporated. For instance, a broadband dual-band operation could be realized by properly notching a rectangular patch. The bandwidth of the higher band for a dual-band PIFA was increased by adding one more resonator. By introducing a stacked element, by making the longer and shorter dipoles resonate, respectively, at slightly below and slightly above the center frequency, or by adding some parasitic structures, the bandwidth at one of the two bands of a dual-band antenna may be increased. Yet, broadband performance is desired at every band of a multi-band antenna.
[0015] Accordingly, there is a need for multi-band broadband antennas. In particular, there is a need for multi-band broadband antennas that are of small size, simple structure, and lightweight materials so as to be easily mounted in a handheld terminal with relatively low cost.
SUMMARY OF THE INVENTIONS
[0016] The inventions satisfy the need for multi-band broadband antennas such as may be used in wireless communication devices. Examples are presented of a broadband planar antenna, of two dual-band antennas, and or a triple-band antenna pursuant to the inventions. The antennas of the inventions have the advantages of being of simple structures such that they may be implemented in a small size, of lightweight materials, and at a relatively low cost.
[0017] The inventions include an antenna made up of two inverted-L antennas (ILAs) facing each other across a gap. This antenna may be referred to as a loop antenna with a gap. One of the ILAs is fed by an input, and may be directly fed by a coaxial cable input. The other ILA is electromagnetically coupled with respect to the fed ILA. The coupled ILA faces the fed ILA, but is separated from the fed ILA by a gap. The length of the coupled ILA is longer than the fed ILA. In particular, the fed ILA, the coupled ILA, and the gap may be positioned with respect to each other to form three sides of a square, and may include a ground plane forming the fourth side of the square. Even more particularly, each of the ILAs may include a vertical leg of the same length that are parallel with respect to each other. Each of the ILAs also may include a horizontal leg, but the horizontal leg of the fed ILA may be shorter than the coupled ILA. In other words, the horizontal leg of the coupled ILA may be longer than the horizontal leg of the fed ILA.
[0018] The inventions also include a dual-band antenna. An exemplary dual-band antenna may include an inverted-L antenna (ILA) referred to as the "first" ILA and another ILA referred to as the "second" ILA. In this example, the second ILA is electromagnetically coupled with respect to the first ILA, faces the first ILA, and is separated from the first ILA by a gap. The second ILA may be longer than the first ILA. In addition to the two ILAs, the exemplary dual-band antenna includes a monopole antenna disposed between the first ILA and the second ILA, and operative to receive input. Further, a connection exists between the monopole antenna and the first ILA to feed input to the first ILA. The connection may connect to the monopole antenna near its base and to the first ILA at its base. Each of the ILAs has a horizontal leg with the horizontal leg of the first ILA being shorter than the horizontal leg of the second ILA. The monopole antenna may be shorter than the vertical leg of the second ILA.
[0019] In addition, the inventions include a triple-band antenna. An exemplary triple-band antenna may include an inverted-L antenna (ILA) referred to as the "first" ILA and another ILA referred to as the "second" ILA. In this example, the second ILA is electromagnetically coupled with respect to the first ILA, faces the first ILA, and is separated from the first ILA by a gap. The second ILA may be longer than the first ILA. In addition to the two ILAs, the exemplary triple-band antenna includes a monopole antenna disposed between the first ILA and the second ILA, and operative to receive input through a feed probe. Further, a connection exists between the monopole antenna and the first ILA to feed input to the first ILA. The connection may connect to the monopole antenna near its base and to the first ILA at its base. A conductor is connected to the monopole antenna opposite to the connection. The conductor extends horizontally from the monopole antenna towards, but not reaching, the second ILA. The conductor and the feed probe combine to form a third ILA in this antenna.
[0020] Further, the inventions include another dual-band antenna. An exemplary dual-band antenna may include an inner cut loop antenna encompassed by an outer cut loop antenna. The inner cut loop antenna may include a "first" inverted-L antenna (ILA) facing a "second" ILA across a "first" gap. The first ILA is fed input while the second ILA is electromagnetically coupled at least to the first ILA. The outer cut loop antenna includes a "third" ILA facing a "fourth" ILA across a "second" gap. The third ILA is fed input via a feed probe and a connection connected to the first ILA of the inner cut loop antenna while the fourth ILA is electromagnetically coupled at least to the third ILA. a
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 illustrates an exemplary loop antenna with a gap for bandwidth enhancement according to the inventions.
FIG. 2 is a graph of the Voltage Standing Wave Ratio (VSWR) for the exemplary antenna of FIG. 1.
FIG. 3 illustrates an exemplary planar dual-band loop-monopole antenna according to the inventions.
FIG. 4 is a graph of the VSWR for the exemplary antenna of FIG. 3.
FIG. 5 illustrates an exemplary planar triple-band loop-monopole antenna according to the inventions
FIG. 6 is a graph of the VSWR for the exemplary antenna of FIG. 5.
FIG. 7 illustrates an exemplary planar dual-band loop-loop antenna according to the inventions.
FIG. 8 is a graph of the VSWR for the exemplary antenna of FIG. 7.
[0028] The inventions include multi-band broadband planar antennas such as may be used with mobile communication devices and in other compact antenna applications. Advantageously, the inventions provide multi-band broadband antennas that may be of small size, simple structure, and lightweight materials so as to be easily mounted in a handheld terminal with relatively low cost.
FIGS. 1-2-Loop Antenna with a Gap
[0030] FIG. 1 illustrates an exemplary broadband planar antenna 10 according to the inventions. In particular, the exemplary broadband planar antenna 10 may be considered a square wire loop antenna on a ground plane 11 with a gap 12, and may be referred to as a loop antenna with a gap. As explained below, the position of the gap 12 in the loop affects the bandwidth of the antenna 10.
[0031] The antenna 10 illustrated in FIG. 1 may also be considered to be comprised of two Inverted-L Antennas (ILAs) 14, 16. In the exemplary embodiment, ILA 14 has a vertical leg 15 of height H connected at its top at a right angle to the right to a horizontal leg 18 of length L1. ILA 14 is directly fed by an input 17 such as a coaxial cable input.
[0032] The other ILA, ILA 16, may be said to face the directly fed ILA 14. ILA 16 has a vertical leg 22 of height H parallel to the vertical leg 15 of ILA 14. ILA 16, like ILA 14, has a horizontal leg 22 connected to the top of its vertical leg 20 at a right angle. But the horizontal leg 22 of ILA 16 is connected at a right angle to the left of its vertical leg 20, and the horizontal leg 22 of ILA 16 is of length L2. In effect, the horizontal leg 18 of ILA 14 faces the horizontal leg 22 of ILA 16 across the gap 12 of the antenna 10. ILA 16 further differs from ILA 14 in that ILA 16 is excited by electromagnetic coupling with respect to the directly fed ILA 14.
[0033] Advantageously, the broadband design of antenna 10 is achieved by making the length of the coupled ILA 16 longer than the directly fed ILA 14. Given that the heights of the vertical legs 15, 20 of the respective ILAs 14, 16 are the same (as noted, the antenna 10 may be considered a square loop antenna with a gap), the longer length of the coupled ILA 16 is achieved by making its horizontal leg 22 longer than the horizontal leg 18 of the directly fed ILA 14. In other words, L2 is greater than L1 as illustrated in FIG. 1.
[0034] The relative lengths of the horizontal legs 18, 22 define the position of the gap 12 in the antenna 10. Thus, a change in the relative lengths causes an adjustment in the position of the gap 12 in the antenna 10. The shorter the horizontal leg 18 of the directly fed ILA 14, the closer the gap 12 in the antenna 10 is to the vertical leg 15 of ILA 14. Conversely, the longer the horizontal leg 18 of the directly fed ILA 14, the closer the gap 12 is to the vertical leg 20 of the coupled ILA 16. The position of the gap 12 affects the bandwidth of the antenna 10.
[0035] FIG. 2 is a graph 24 of frequency (GHz) vs. simulated Voltage Standing Wave Ratio (VSWR) for the exemplary antenna 10 of FIG. 1 with different gap positions. The simulation was carried out using the MoM (Method of Moment) based Numerical Electromagnetics Code (NEC V1.1) and under the assumption of an infinite ground plane 11. Graph 24 includes a table 26 with three entries relating to the respective lengths of the horizontal legs 18, 22 of the ILAs 14, 16 used in the simulation. Each entry includes a measured length of the horizontal leg 18 of the directly fed ILA 14 and a measured length of the horizontal leg 22 of the coupled ILA 16. Each entry relates to the simulation and is plotted on the graph 24. Note, in this example, the gap 12=2 mm.
[0036] FIG. 2 illustrates that as the difference between the length L2 of the horizontal leg 22 of the coupled ILA 16 and the length L1 of the horizontal leg 18 of the directly fed ILA 14 (e.g., L2-L1) decreases, the respective resonant frequencies for the ILAs 14, 16 (FHI for ILA 14 and FLO for ILA 16) move closer to each other. The maximum bandwidth for a certain criterion of VSWR is obtained when all the VSWR within this frequency band is below the VSWR threshold. For this example, the bandwidth for a VSWR criterion=2 is calculated to be 35%. Therefore, the optimum VSWR of 2 or less is achieved for a very wide bandwidth.
FIGS. 3-4-Dual-Band Antenna
[0038] FIG. 3 illustrates an exemplary dual-band broadband planar antenna 30 according to the inventions. The antenna 30 of FIG. 3 is similar to the antenna 10 of FIG. 1 in that each may be considered a square wire loop antenna on a ground plane 11 with a gap 12. The antenna 30 of FIG. 3 differs and provides dual-band operation by the addition of a monopole antenna 32 in the middle of the antenna 30 plus some adjustments. A monopole antenna may also be referred to as a monopole herein.
[0039] More particularly, like the antenna 10 of FIG. 1, the antenna 30 of FIG. 3 may be considered to be comprised of two Inverted-L antennas (ILAs) 34, 36 that face each other across a gap 12. One of the ILAs 34 is fed input (as explained below), and the other ILA 36 is electromagnetically coupled to the fed ILA 34 and/or coupled with respect to the other parts of the antenna 30. Each of the ILAs 34, 36 includes a vertical leg, respectively 35, 40.
[0040] The antenna 30, however, differs from the antenna 10 because the antenna 30 has a vertical monopole 32 rising from the ground plane 11 and centered between vertical legs 25, 30 of the ILAs 34, 36 of the antenna 30. The monopole 32 has a length less than the length (or height) of the vertical legs 25, 30 of the ILAs 34, 36. The monopole 32 is fed from an input 33, such as by a coaxial cable input, which also feeds ILA 34 through a connection 37 from the monopole 32 to the vertical leg 35 of the ILA 34. For example, as illustrated in FIG. 3, the input 33 may be centered between the vertical legs 35, 40 of the ILAs 34, 35 to directly feed the monopole 32 and to feed the ILA 34 through the connection 37 between the monopole 32 and the vertical leg 35 of the ILA 34.
[0041] In particular, the connection 37 is disposed between the monopole 32 and the leg 35 of the fed ILA 34 such that the connection 37 connects near the base or input end of the monopole 32, runs above and parallel to the ground plane 11, and connects to the end closest to the ground plane 11 of the vertical leg 35 of the fed ILA 34. Thus, the fed ILA 34 does not connect to the ground plane 11 in antenna 30. As illustrated in FIG. 3, the distance between the ground plane 11 and the connection 37 is h1, which may also be referred to as the height of the connection 37. The length of the vertical leg 35 of ILA 34 is H2. The length of the vertical leg 40 of the coupled ILA 36 is h1+H2.
[0042] The introduction of the monopole 32 as part of the antenna 30 causes additional differences with respect to the antenna 10 of FIG. 1. For example, the fed ILA 34 of antenna 30 includes a horizontal leg 38 of length L3. The coupled ILA 36 of antenna 30 includes a horizontal leg 42 of length L4. The respective lengths of L3 and L4 may need adjustment (as compared to their analogous parts in antenna 10) due to the connection 37. The monopole 32 is designed for resonance at a higher frequency than the ILAs. The height (h1) of the connection 37 is optimized for an optimal VSWR. Note that the connection 37 (which may be a wire) has a negligible contribution to the radiation fields due to its proximity (h<<H2) to the ground plane 11 (the radiation fields from the connection 37 will be cancelled by its image below the ground plane). This is the reason why only a slight adjustment may be needed for the position of the gap 12.
[0043] FIG. 4 is a graph 44 of frequency (GHz) vs. simulated Voltage Standing Wave Ratio (VSWR) for the exemplary antenna 30 of FIG. 3. The graph 44 illustrates the calculated VSWR for a dual-band operation in 1 GHz and 2 GHz bands where L3=12 mm; L4=36 mm; H2=46 mm; h1=4 mm; the gap 12=2 mm; the monopole=41 mm (from the connection 37 to the end of the monopole opposite the ground plane); and the wire radius=1 mm.
[0044] Graph 44 illustrates there are two distinct bandwidths where the VSWR is less than 2: a lower area 46 and an upper area 48. Advantageously, the upper area 48 stretches over a wide band of frequencies. The VSWR in the upper area (or higher band) 48 is quite low and has a flat variation (VSWR<=1.5 from 1.6 to 2.5 GHz). Such a dual and broadband antenna is suitable for use in AMPS/PCS, GSM/DCS, PDC/PHS, IMT2000 and 2.4 GHz ISM band WLAN.
FIGS. 5-6-Triple-Band Antenna
[0046] FIG. 5 illustrates an exemplary triple-band broadband planar antenna 50 according to the inventions. A triple-band antenna may be particularly advantageous so as to be used in connection with the 5 GHz ISM band for WLAN applications in mobile devices and other units.
[0047] The antenna 50 of FIG. 5 is similar to the antenna 30 of FIG. 3, but for the addition of a wire (also referred to as conductor) 51 that is connected to the monopole antenna 52 opposite to the connection 57 between the monopole antenna 52 and the vertical leg 55 of the ILA 54. The addition of the conductor 51 allows for triple band operation of the antenna 50.
[0048] Particularly, the antenna 50 of FIG. 5 may be considered to be comprised of two Inverted-L antennas (ILAs) 54, 56 that face each other across a gap 12. ILA 54 includes a vertical leg 55 and horizontal leg 58, which is of length L5. ILA 56 includes a vertical leg 60 and a horizontal leg 62, which is of length L6.
[0049] A vertical monopole antenna 52 is disposed between the ILAs 54, 56. The monopole 52 is fed through a feed probe 59 from an input 53, which also feeds ILA 54 through a connection 57 from the monopole 52 to the vertical leg 55 of the ILA 54. The connection 57 connects near the base or input end of the monopole 52, runs above and parallel to the ground plane 11, and connects to the end closes to the ground plane 11 of the vertical leg 55 of the fed ILA 54. As illustrated in FIG. 5, the distance between the ground plane 11 and the connection 57 is h2. In the exemplary embodiment, the feed probe 59 between the input 53 has the height of h2. The length of the vertical leg 55 of ILA 54 is H3. The length of the vertical leg 60 of ILA 56 is h2+H3. ILA 56 is electromagnetically coupled to ILA 54 and/or may be coupled to the other parts of the antenna 50.
[0050] As noted, a wire or conductor 51 is connected to the monopole antenna 52 opposite to the connection 57. The conductor 51 extends horizontally from the monopole 52 in the direction of, but does not reach, the vertical leg 60 of the ILA 56. The conductor 51 with the feed probe 59 acts as an ILA and allows for three band operation of antenna 50. In the example described in connection with FIGS. 5 and 6, the ILA composed of the conductor 51 and the feed probe 59 acts with respect to the 5 GHz band. Given its configuration including the 2 ILAs 54, 56 forming a loop (but for the gap 12), the monopole 52, and the ILA composed of the conductor 51 and the feed probe 59, the antenna 50 may be referred to as a triple-band loop-monopole-ILA. Note that the radiation contribution from the connection 57 and/or the conductor 51 is no longer negligible in the 5 GHz band since h2 becomes comparable to a fraction of one wavelength in this example.
[0051] FIG. 6 is a graph 64 of frequency (GHz) vs. simulated Voltage Standing Wave Ratio (VSWR) for the exemplary antenna 50 of FIG. 5. The graph 64 illustrates the calculated VSWR for a triple-band operation where L5=12 mm; L6=36 mm; H3=46 mm; the gap=2 mm; the monopole 52=10 mm; the conductor 51=10 mm; and the wire radius=1 mm.
[0052] Advantageously, a third, additional broadband (38%) is obtained in the 5 GHz band (or band 3) over the previous exemplary antenna 30 described in connection with FIGS. 3-4. This broadband performance also benefits from a combination of the fundamental mode of the additional ILA (the conductor 51 and the feed probe 59) and the high-order modes of the two ILAs 54, 56 and the monopole 52. The addition of the ILA (the conductor 51 and the feed probe 59) does not affect the broadband performance of the original dual-band antenna (antenna 30) in the lower 1 GHz and 2 GHz bands.
FIGS. 7-8-Dual-Band Loop-Loop Antenna
[0054] FIG. 7 illustrates another exemplary dual-band broadband planar antenna 70 according to the inventions. In some applications, an antenna may only need to cover the 2 GHz and 5 GHz bands. In such circumstances, the physical size of the antenna may be reduced, but there is a need to increase the bandwidth of the lower band in order to cover all the mobile communication and WLAN applications in the 2 GHz band. This need can be satisfied through an introduction of two cut loops, which results in a dual-band loop-loop antenna. An example of such an antenna is shown in FIG. 7.
[0055] The exemplary antenna 70 of FIG. 7 includes an inner cut loop 71 and an outer cut loop 72. As the terms imply, the inner cut loop 71 is set within the outer cut loop 72. The inner cut loop 71 includes two ILAs 73, 74, which are positioned with respect to each other (like in the previously described antenna examples) so that the ILAs face each other across a gap 75. The outer cut loop 72 also includes two ILAs 76, 77, which are also positioned so that the ILAs face each other across a gap 78.
[0056] Both the inner cut loop 71 and the outer cut loop 72 include an ILA that is fed input 79 with the other ILA in the loop being electromagnetically coupled. With respect to the inner cut loop 71, the ILA 73 is directly fed while the ILA 74 is electromagnetically coupled. With respect to the outer cut loop 72, the ILA 77 is fed from input 79 via feed probe 80 and connection 81. The configuration of the feeding of ILA 77 is similar to the feeding of ILA 54 as described in connection with antenna 50 shown in FIG. 5.
[0057] Further, the coupled ILA 74 of the inner cut loop 71 has a vertical leg 82 of height H5 and a horizontal leg 83 of L10. The fed ILA 73 of the inner cut loop 71 has a vertical leg 84 whose height, when combined with the height of the feed probe 80, equals the height of the vertical leg 82 of the coupled ILA 74. The fed ILA 73 also has a horizontal leg 85 of length L9.
[0058] The fed ILA 77 of the outer cut loop 72 has a vertical leg 86 of a height H4. The fed ILA 77 also has a horizontal leg 87 of length L7, which is also the length of the connector 81. The coupled ILA 76 of the outer cut loop 72 has a vertical leg of a height H4+h3 where h3 is the height of the connector 81 between the fed ILA 73 of the inner cut loop 71 and the fed ILA 77 of the outer cut loop 72. The coupled ILA 76 has a horizontal leg of length L8.
[0059] The simulated VSWR of the exemplary dual-band loop-loop antenna 70 is plotted in the graph 94 shown in FIG. 8. The bandwidth of the lower band is increased to 44% from 31% and the bandwidth of the higher band keeps 55%. The increase in the bandwidth in the lower band (band 1) is attributed to the combination of three resonant frequencies, which respectively correspond to three ILAs: the fed ILA 77 of the outer cut loop 72; the coupled ILA 76 of the outer cut loop 72; and the coupled ILA 74 of the inner cut loop 71. The fed ILA 73 of the inner cut loop 71 has a similar function in the antenna 70 shown in FIG. 7 as the monopole antenna 52 in FIG. 5, which leads to a broadband performance in the higher band (band 2).
CONCLUSION
[0060] Advantageously, the features and functions of the inventions described herein allow for their use in many different manufacturing configurations. For applications in a wireless communication handheld terminal (e.g., a mobile phone handset), an antenna per the inventions can be printed on a printed circuit board (PCB) or an electrically thin dielectric substrate (e.g. RT/duroid 5880). The printed piece can be mounted either (a) at the top of the handset backside or (b) at the bottom of the front side of the handset. The top-mounted configuration can serve as a "flip" cover of the handset while the bottom-mounted mouthpiece can be integrated with a microphone.
[0061] From the foregoing description of the exemplary embodiments of the inventions and operation thereof, other embodiments will suggest themselves to those skilled in the art. Therefore, the scope of the inventions is to be limited only by the claims below and equivalents thereof.
US 2003107518
Folded shorted patch antenna
Folded shorted patch antenna
Inventor(s): LI RONGLIN [US]; LASKAR JOY [US]; TENTZERIS EMMANOUIL [US] + (LI RONGLIN, ; LASKAR JOY, ; TENTZERIS EMMANOUIL)
Applicant(s): LI RONGLIN, ; LASKAR JOY, ; TENTZERIS EMMANOUIL
Classification: - international: H01Q1/24; H01Q5/00; H01Q9/04; (IPC1-7): H01Q1/24 - European: H01Q1/24A1A; H01Q5/00C; H01Q9/04B1; H01Q9/04B2
Abstract -- A patch antenna is described that includes a ground plane, a first shorting structure in contact with the ground plane, a first conductor plate in contact with the first shorting structure. The patch antenna can also include a second shorting structure in contact with the ground plane, and a second conductor plate in contact with the second shorting structure and forming a radiation slot with the first conductor plate. Other devices and methods are herein provided for.
TECHNICAL FIELD
[0002] The present invention is generally related to communications, and, more particularly, is related to antennas.
BACKGROUND OF THE INVENTION
[0003] In modern mobile and wireless communications systems, there is an increasing demand for smaller low-cost antennas. This is especially true for handheld wireless applications, such as in mobile phone handsets or Bluetooth chips, where a package-integrated antenna may be desirable. It is well known that planar structures such as microstrip patch antennas have a significant number of advantages over conventional antennas, such as low profile, light weight and low production cost. However, in some practical wireless communications systems such as Global System for Mobile Communications (GSM) 1800, Personal Communications Service (PCS) 1900, wideband code division multiple access standard IMT 2000, or Bluetooth ISM (Industrial, Scientific, and Medical), the physical size of planar structures may be too large for integration with radio frequency (RF) devices.
[0004] One type of antenna suitable for use with personal communications devices is the conventional patch antenna 100, shown in a side view in FIG. 1. The patch antenna 100 (here a [lambda]0/2 patch antenna) comprises a ground plane 102, a patch (or a conductor plate) 104, and a feed 106. It is well known that a conventional patch antenna operating at the fundamental mode, Transverse Magnetic (TM) mode TM01, has an antenna length of [lambda]0/2. The length of the patch is set in relation to a wavelength [lambda]0 associated with the resonant frequency f0. A number of techniques have been proposed to reduce the size of conventional half-wave ([lambda]0/2, where [lambda]0 is the guide wavelength in the substrate) patch antennas. One approach is to use a high dielectric constant substrate (e.g., between the patch 104 and the ground plane 102). However, such an approach often leads to poor efficiency and narrow bandwidth.
[0005] Shorting structures (e.g., shorting posts, shorting walls) also have been used in different arrangements to reduce the overall size of the patch antenna. Considering that the electric field is zero for the TM01 mode at the middle of the patch 104, the patch 104 along its middle line can be shorted with a metal wall without significantly changing the resonant frequency of the patch antenna 100. FIG. 2 illustrates a conventional shorted patch antenna 200 that includes a patch 204 that is shorted to the ground plane 202 with a metal wall 208. This shorted patch antenna 200 includes a patch 204 with a length of [lambda]0/4. Further patch size reduction measures include using a shorting pin (not shown) near the feed 206. The size-reduction technique using a shorting pin has been applied to the design of small patch antennas for 3G IMT-2000 mobile handsets.
[0006] A planar invert-F antenna (PIFA) is one of the most well-known and documented small patch antennas. Actually, the PIFA can be viewed as a shorted-patch antenna. Therefore the antenna length of a PIFA is generally less than [lambda]0/4. When a shorting post is located at a corner of a square plate, the length of the PIFA can be reduced to [lambda]0/8. The size of a PIFA can be also reduced by loading it. Recent research efforts on the size reduction of patch antennas have focused on patch-shape optimization to increase the effective electric length of the patch. For example, by notching a rectangular patch, the antenna length can be reduced to less than [lambda]0/8. A printed antenna with a surface area 75% smaller than a conventional microstrip patch was obtained by incorporating strategically positioned notches near a shorting pin. However, the demand for a further reduction in size while preserving or improving some performance characteristics of larger antennas still exists.
[0007] Thus, a need exists in the industry to address the aforementioned and/or other deficiencies and inadequacies.
SUMMARY OF THE INVENTION
[0008] The preferred embodiments of the present invention provide for a patch antenna. Briefly described, one embodiment of the patch antenna, among others, can be implemented as follows. The patch antenna includes a ground plane, a first shorting structure in contact with the ground plane, a first conductor plate in contact with the first shorting structure, a second shorting structure in contact with the ground plane, and a second conductor plate in contact with the second shorting structure and forming a radiation slot with the first conductor plate.
[0009] The preferred embodiments of the present invention also include, among others, a method for making a patch antenna. One method can generally be described by the following steps: connecting a first conductor plate to a ground plane with a first shorting structure, the first conductor plate substantially parallel to the ground plane, the first conductor plate having an electrical length of approximately [lambda]0/16; and connecting a second conductor plate to the ground plane with a second shorting structure, the second conductor plate substantially parallel to the first conductor plate, the second conductor plate having an electrical length of approximately [lambda]0/16, the second conductor plate forming a radiation slot with the first conductor plate.
[0010] Other systems, methods, features, and advantages of the present invention will be or become apparent to one with skill in the art upon examination of the following drawings and detailed description. It is intended that all such additional systems, methods, features, and advantages be included within this description, be within the scope of the present invention, and be protected by the accompanying claims.
BRIEF DESCRIPTION OF THE DRAWINGS
[0011] Many aspects of the invention can be better understood with reference to the following drawings. The components in the drawings are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the present invention. Moreover, in the drawings, like reference numerals designate corresponding parts throughout the several views.
FIG. 1 is a side view of a prior art patch antenna.
FIGS. 3A-3B are front and rear view schematic diagrams of a portable telephone that incorporates a folded shorted patch (FSP) antenna, in accordance with one embodiment of the invention.
FIG. 5A is an isometric view of the FSP antenna depicted in FIG. 4B, in accordance with one embodiment of the invention.
FIG. 5B is a Smith chart showing the input impedance of the FSP antenna of FIG. 5A fed at different lower patch locations, in accordance with one embodiment of the invention.
FIGS. 6-8 are graphs showing the effect on return loss and resonant frequency when modifying the shape parameters of the FSP antenna of FIG. 5A, in accordance with one embodiment of the invention.
FIGS. 9A-9B are graphs showing the radiation patterns of the FSP antenna of FIG. 5A after modifying the height parameters, in accordance with one embodiment of the invention.
FIGS. 10A-10C are side views illustrating the process of unfolding a folded shorted patch (S-P) antenna to arrive at a transmission model, in accordance with one embodiment of the invention.
FIGS. 11A-11C are Smith charts comparing the theoretical and numerical input impedance of the unfolded S-P antennas and folded S-P antennas depicted in FIGS. 10A-10C, in accordance with one embodiment of the invention.
FIG. 12 is a graphical illustration of the suseptance and capacitance versus various resonant frequencies of the unfolded S-P antennas and folded S-P antennas depicted in FIGS. 10A-10C, in accordance with one embodiment of the invention.
FIG. 13 is a graph showing the simulated results for input impedance versus frequency for the FSP antenna using a lumped capacitor, in accordance with an alternate embodiment of the invention.
FIG. 14 is a graph showing the difference between simulated and measured return loss versus resonance frequency for one example FSP antenna implementation, in accordance with one embodiment of the invention.
FIGS. 15A-15B are graphs showing the radiation patterns of the simulated versus measured results of the FSP implementation described in association with FIG. 14, in accordance with one embodiment of the invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0027] The preferred embodiments of the invention now will be described more fully hereinafter with reference to the accompanying drawings. One way of understanding the preferred embodiments of the invention includes viewing them within the context of a personal communications device, and more particularly within the context of an antenna for a portable telephone. However, it is noted that the preferred embodiments can be viewed within other contexts, such as for use in cellular handsets, sensors for monitoring, and wireless smart cards, among other example contexts that use antennas for transmitting and/or receiving signals over a medium.
[0028] In the description that follows, a folded shorted patch (FSP) antenna will be described that is reduced in size compared to conventional patch antennas. By folding a shorted rectangular patch, the resonant length of the antenna can be reduced from [lambda]0/4 to [lambda]0/8. A further decrease of as much as more than 50% in the resonant length may be achieved through adjusting the width of the shorting walls and the heights of the folded patches. Thus the overall electrical length (less than [lambda]0/16) of the FSP antenna can be eight times shorter than the length of a conventional patch ([lambda]0/2). A brief note about the term electrical length can be described as follows. For example, if a patch with a physical length of 150 millimeters (mm) can operate at 1 gigahertz (GHz) ([lambda]0=300 mm), then the electrical length of this patch will be understood to be [lambda]0/2. But if the patch with the same physical length (150 mm) can operate at 500 megahertz (MHz) ([lambda]0=600 mm), the electrical length of the same patch is now [lambda]0/4.
[0029] A structure of the FSP antenna for a personal communications device will be described below. One method for making the FSP antenna will also be described, as well as some numerical simulations described that are recorded in a series of graphs illustrating input impedance, radiation patterns, and the effect on return loss and resonant frequency when various elements of the FSP antenna are modified. This discussion is followed by a theoretical analysis based on a transmission-line model created by unfolding a folded shorted patch antenna, and then a comparison of the theoretical versus numerical simulations is discussed and illustrated. The FSP antenna operation for reducing resonant frequency is analyzed by considering the antenna as a shorted patch loaded with a capacitive device, followed by an example implementation of an FSP antenna.
[0030] FIGS. 3A and 3B illustrate one example implementation for the FSP antenna. Specifically, FIG. 3A depicts a front view of a portable phone 300 having a speaker 308, a microphone 312, a display 316, and a keyboard 320, as well as internal transceiver circuitry not shown. FIG. 3B is a rear view of the portable phone 300 shown in FIG. 3A showing an FSP antenna 504 preferably mounted to the back of the portable phone 300 to reduce the specific absorption rate (SAR) potentially absorbed in the head of a user. The length of the FSP antenna 504 determines its resonant frequency. For example, a quarter wave (i.e., [lambda]0/4) patch antenna having a length L will resonate at a frequency of c/4L, where c equals the speed of light. At or near the resonant frequency is where the FSP antenna 504, or patch antennas in general, radiate most effectively.
[0031] FIGS. 4A-4B show a series of side views demonstrating one mechanism for making the FSP antenna structure via a series of folding operations, in accordance with one embodiment of the invention. FIG. 4A shows a folded shorted patch antenna 400 that demonstrates the steps of folding over the patch 404 together with the ground plane 402. The example folded shorted patch antenna 400 includes a lower shorting wall 408 and a feed probe 406. The total resonant length of the folded shorted patch antenna 400 is still [lambda]0/4. That is, the length spanning from the shorting wall formed by folding the ground plane 402 (referenced as the upper shorting wall 510 in FIG. 4B) to the radiating slot entrance is [lambda]0/4, which indicates that the resonant frequency of an FSP antenna 504 (FIG. 4B) is similar to that of a conventional shorted patch antenna 200 (FIG. 2), as is borne out in numerical simulations and theoretical analysis. The actual length (i.e., electrical length) of the folded patch 404 has been reduced through the folding operation by 50% to [lambda]0/8.
[0032] With continued reference to FIG. 4A, and referring now to FIG. 4B, by adding a new piece of the ground plane to the right of the folded ground plane 402 and pressing the folded patch 404 together to form a lower patch 505, a folded shorted patch antenna 504 is produced. Note that the original right part of the folded ground plane 402 (FIG. 4A) now serves as an upper shorting wall 510 and an upper patch 512 of the folded shorted patch antenna 504. The space between the upper patch 512 and the lower patch 505 comprise a radiating slot from which electromagnetic energy is concentrated and transmitted and/or received.
[0033] FIG. 5A depicts a general structure of the FSP antenna 504 shown in FIG. 4B. For simplicity, the discussions that follow will assume an implementation for the FSP antenna 504 in free space (i.e., an air dielectric substrate is approximated as a free space). The FSP antenna 504 includes a ground plane 502, a lower patch 505, an upper patch 512, a lower shorting wall 508, an upper shorting wall 510, and a feed probe 506. The ground plane 502 is preferably made of a conductive material such as aluminum, copper, and/or gold. The ground plane 502 is separated from the lower patch 505 by a dielectric substrate. The dielectric substrate described herein will be air, but can be glass or practically any other dielectric substrate.
[0034] The lower patch 505 is approximately parallel to the ground plane 502, and is shown with dimensions of width W1, length L1, and a height h1 from the ground plane 502. One end of the lower patch 505 is in contact with the ground plane 502 via the lower shorting wall 508. The lower shorting wall 508 is shown with dimensions of width d1.
[0035] A feed probe 506 can be electrically connected to the lower patch 505. The feed probe 506, which can be a coaxial cable, passes through the ground plane 502 and contacts the lower patch 505. For example, a coaxial cable having an inner and outer conductor will be connected to the lower patch 505 using the inner conductor (e.g., feed probe, with no connection to the ground plane) and the outer conductor will connect to the ground plane 502. The feed probe 506 connects a signal unit (not shown) to the lower patch 505 at various distances (yp) from the lower shorting wall 508 in the y-direction. The signal unit can be connected to the lower patch 505 in other ways, such as via a microstrip or a transmission line. The signal unit provides a signal of a selected frequency band to the lower patch 505, which creates a surface current in the lower patch 505. The density of the surface current is high near the region of the lower patch 505 in proximity to where the feed probe 506 contacts the lower patch 505. This current density decreases gradually along the length of the lower patch 505 in a direction away from the point where the feed probe 506 contacts the lower patch 505.
[0036] The FSP antenna 504 can be adjusted to match a defined feed input impedance, for example a 50-[Omega] feed, by changing the position of the feed probe 506. The input impedance of the FSP antenna fed at different positions (yp) is plotted in a Smith chart shown in FIG. 5B, with position adjustment in the x-direction having little effect on the impedance match. As shown, the impedance locus shrinks in size as the feed point moves closer to the lower shorting wall 508 (FIG. 5A). The asymmetry of the impedance locus about the x=0 axis in the Smith chart is due to the feed-probe reactance, which when read from the impedance locus is found to be near j25 [Omega].
[0037] Returning to FIG. 5A, the FSP antenna 504 also includes an upper patch 512 that is approximately parallel to the lower patch 505. The upper patch 512 serves as a coupling patch (i.e., it is not fed by direct physical contact to a feed line or feed probe, but instead is excited through electromagnetic coupling). The upper patch 512 is shown with dimensions of width W2, length L2, and a height h2 from the lower patch 505. The upper patch 512 is in contact with the ground plane 502 via the upper shorting wall 510. The upper shorting wall 510 is shown with a width of d2. The electric field of the FSP antenna 504 is concentrated in the gap (i.e., radiation slot) between the lower and upper patches (505, 512). Surface-current distributions primarily occur on the top face of the lower patch 505, with smaller surface current distributions occurring on the inside face of the upper shorting wall 510. An electric-field concentration also exists between the edge of the lower patch 505 (the edge closest to the upper shorting wall 510) and the upper shorting wall 510. This is due at least in part to the effects of the relatively sharp edge of the lower patch 505 and the short distance between the edge and the upper shorting wall 510. Increasing the distance between the edge and the upper shorting wall 510 (i.e., a shortened L1) can improve the impedance bandwidth of the FSP antenna 504.
[0038] With continued reference to FIG. 5A throughout the discussion of FIGS. 6-8 that follow, the resonant frequency of the FSP antenna 504 can be lowered by slightly modifying the shape parameters of the FSP antenna 504, such as by reducing the widths of the two shorting walls 508 and 510 and/or adjusting the heights h1, h2 of the lower and upper patches 505, 512. FIGS. 6-8 provide illustrations of the effects on return loss and resonant frequency when simulating the modification of these dimensions through numerical analysis (e.g., via well-known transmission line match (TLM) and finite differential time domain (FDTD) simulations). FIG. 6 shows the simulated effects on resonant frequency and return loss with a varying d1 dimension. For example, the width (d1) of the lower shorting wall 508 is reduced while setting and maintaining the width (d2) of the upper shorting wall 510 to be d2=W2 and the heights (h1=h2=1.5 millimeters (mm)) of the lower and upper patches 505, 512. As shown, the resonant frequency (shown at the inverted peaks) decreases as the width (d1) of the lower shorting wall 508 becomes narrower (i.e., from 10 mm to 2 mm). Continuing the analysis, while setting and maintaining d1=2 mm, the width of the upper shorting wall (d2) can be changed, the effect of which is shown in FIG. 7. Again, the resonant frequency further decreases as d2 reduces. One reason for the decrease of the resonant frequency with a reduction of the widths of the shorting walls (508, 510) is an increase in the inductance of the upper and lower patches (505, 512).
[0039] FIG. 8 demonstrates the effects of simulating an adjustment in the height (h1) of the lower patch 505 while setting and maintaining d1=d2=2 mm and the total FSP antenna height (h1+h2)=3 mm. The variation of the return loss with h1 and the difference in resonance frequency is as shown. It is noted that a variation in h1 has a more significant impact on the resonant frequency than changes in d1 and d2. As the lower patch 505 moves toward the upper patch 512, the resonant frequency decreases. When the distance between the lower and upper patches (505, 512) is less than [1/5] of the total FSP antenna height, the resonant frequency reduces by more than a half of 3.6 GHz. One reason for the decrease in the resonant frequency with increase in h1 (or a decrease in the distance between the lower and upper patches (505, 512)) is due to an enhancement of the capacitive coupling between the lower and upper patches (505, 512) as the upper and lower patches are brought closer to each other.
[0040] The position of the feed probe 506 will typically be adjusted for different antenna shape parameters to match, for example, a 50-[Omega] feed. Usually the radiation resistance increases with a decrease in antenna thickness and patch width because the radiated power decreases. Thus, the resonant resistance increases as the resonant frequency decreases. For the FSP antenna 504, the more the resonant frequency is reduced by varying the antenna shape parameters, the closer the feed probe position is shifted to the lower shorting wall 508.
[0041] The simulated radiation patterns at resonant frequencies for h1=0.5 mm at 3.63 GHz and with h1=2.5 mm at 1.65 GHz are shown in FIGS. 9A and 9B. As shown in FIG. 9A, the radiation pattern represents the far-zone field in the x-z plane of a Cartesian coordinate system (x,y,z) while FIG. 9B includes a radiation pattern that represents the far-zone field in the y-z plane. In each plane, the far-zone field includes two orthogonal components E[phi] and E[theta]. E[phi] in the y-z plane is zero due to symmetry, and thus there are only two lines indicated in FIG. 9B. For comparison, the radiation patterns at two different frequencies are plotted in each graph. The radiation patterns for the h1=0.5 mm case is depicted using a solid line, and the h1=2.5 mm case is depicted with a dotted line. The magnitude of electromagnetic energy, ¦E¦, is in units of decibels (dB). The cross-polarized component is shown in FIG. 9A, and illustrates a more pronounced difference between the two cases: a lower h1 corresponds to a higher cross-polarized level. Usually the cross polarized level increases with antenna thickness (i.e., total antenna height). When h1 decreases, h2 increases and the resonant frequency increases. As a result, the width of the radiating slot (h2) further increases electrically, thus causing an increase in the cross-polarized level.
[0042] In the section that follows, the FSP antenna 504 (FIG. 5A) is described analytically by employing a transmission-line model. Also a qualitative analysis of the resonant frequency of the FSP antenna 504 is presented of the FSP antenna operation.
[0043] FIGS. 10A-10C present the FSP antenna 504 with three different patch-height arrangements, shown in FIGS. 10A-10C under the column heading, "folded S-P" (shorted patch): Case I (h1=h2=1.0 mm), Case II (h1=0.5 mm, h2=1.0 mm), and Case III (h1=1.0 mm, h2=0.5 mm). The "folded S-P" is unfolded to arrive at an "equivalent" (i.e., equivalent for transmission line analysis purposes) unfolded shorted patch (under the column heading, "unfolded S-P") configuration associated with these three cases. Neglecting the effect of discontinuities, the "unfolded S-P" can be represented by a transmission-line equivalent circuit as shown in FIG. 10D. The input impedance of the "unfolded S-P" based on this equivalent circuit is obtained as follows:
Zin=jXf+Z1 (1)
[0044] where Xf is the feed-probe reactance given by
EMI1.1
[0045] with [beta]=2[pi]/[lambda]0 and rp=the feed-probe radius. Z1 (=1/Y1) is obtained from the transmission-line equivalent circuit, that is,
EMI2.1
[0046] where Y01 and Y02 are respectively the characteristic admittance of the lower and upper patches, and Ys=Gs+jBs. Here, Gs is the conductance associated with the power radiated from the radiating edge (or the radiating slot), and Bs is the susceptance due to the energy stored in the fringing field near the edge of the patch. In the calculations described herein, the following equations for Y(=Y01 for h=h1 or Y02 for h=h2), Gs, and Bs were used:
EMI3.1
Bs=Y02 tan([beta][Delta]l) (7)
EMI4.1
[0047] where W is the width of the patch and coefficients [zeta]1, [zeta]3, [zeta]4, [zeta]5 can be found in the reference entitled, "Microstrip antenna design handbook", by R. Garg et al., 2001, which is herein incorporated by reference.
[0048] The theoretical results for the input impedance are obtained using the above analytical expressions and compared in FIGS. 11A-11C with numerical simulations for the above three cases. Note that the numerical results are obtained for the "folded S-P" shown in FIGS. 10A-10C. The theoretical and numerical results are in good agreement. The difference between the theoretical and simulated resonant frequencies is less than 3%. Also, it is again noted that the resonant frequency decreases as h2/h1 decreases. This can be explained qualitatively as follows. For simplicity, the effects of YS(YS<<Y0 in practice) and Xf (focusing on the resonance of the patch alone) are neglected. As a result the "unfolded S-P" becomes a shorted transmission line loaded with an open transmission line. Assume that the resonant frequency is almost independent of the feeding position, yp=L1 Thus, Y1 becomes
EMI5.1
[0049] At resonance, Y1=0 leads to
Y01/tan([beta]L1)=Y02 tan([beta]L1) or tan([beta]L1)={square root}{square root over (Y01/Y02)} (10)
[0050] From equation 5 above, note that Y0 is inversely proportional to h; therefore, from equation 10, it is determined that the resonant frequency varies proportionally with h2/h1. A graphical solution of equation 10 for resonant frequency is depicted in FIG. 12, where the intersection of the curves Y01/tan([beta]L1) and Y02 tan([beta]L1) implies a resonant point. FIG. 12 includes a plot of suseptance versus [beta]L1. Note that if Y01=Y02, then [beta]L1=[pi]/4 corresponds to an antenna length of L1=[lambda]0/8. Also note that an increase in Y02 leads to a decrease in [beta]L1 if Y01 remains unchanged.
[0051] With continued reference to FIGS. 10A-10C, considering the upper patch as a capacitive load provides additional insight for the above analysis. Replacing the upper patch with a capacitor C (not shown), which is connected between the radiating edge of the lower patch and the ground plane of the folded S-P antenna shown in FIGS. 10A-10C, equation 9 becomes
Y01/tan([beta]L1)=[omega]C. (11)
[0052] A graphical solution of equation 11 is also plotted in FIG. 12. As noted, the resonant frequency increases as the capacitance C increases. The resonant length of a capacitively loaded shorted patch will reduce to L1=[lambda]0/8 if the loaded capacitance is C=Y01/[omega]0, where [omega]0=3[pi]/(4L1)*10<8 >rad-s<-1 >is obtained from [beta]L1=/4[pi]. A decrease in h2 is equivalent to an increase in the coupling capacitance between the upper and lower patches, thus eventually leading to a decrease in the resonant frequency.
[0053] Equation 11 suggests an alternate embodiment for the FSP antenna 504 (FIG. 5A), wherein the resonant frequency can be reduced using a lumped capacitive load (e.g., a lumped capacitor between the radiating edge of the lower patch 505 and the ground plane 502 of the FSP antenna 504 of FIG. 5A, as described above). The simulated results for input impedance versus frequency are shown in FIG. 13, wherein the resistance is shown with a sold line and the reactance is shown with a dashed line. As expected, the resonant frequency decreases with an increase in the loaded capacitance. Comparing FIGS. 12 and 13, it is noted that the proportional relationship of the resonant frequencies among C=0.3, 0.6, and 1.2 picofarad (pf) is very similar to that (about 3:4:5) read from the graphical solutions of equation 11 when C=(Y01/[omega]0)/2, C=Y01/[omega]0, and C=2Y01/[omega]0. This demonstrates agreement between the numerical investigation and theoretical analysis described above.
[0054] As one example implementation, a test FSP antenna was integrated in the package of a Bluetooth chip operating in the Bluetooth ISM band (2.4-2.483 GHz). The test FSP antenna was fabricated with a brass sheet with a thickness of 0.254 mm. The following FSP antenna dimensions were chosen: 15 mm*15 mm ( [lambda]0/8*[lambda]0/8). To achieve the bandwidth (near 4%) required by the Bluetooth specifications, the total thickness of the antenna was selected to be 6 mm. By adjusting the height (h1) of the lower patch to 2.85 mm, the resonant frequency can be tuned to approximately 2.44 GHz. The simulated and measured results for the return loss are plotted in FIG. 14. As shown, good performance agreement is obtained, and both of the simulated and measured 10-dB return-loss bandwidths cover the Bluetooth band. The radiation patterns simulated and measured in the xz- and yz-planes at 2.44 GHz were compared, as shown in FIGS. 15A-15B, and good agreement was again noted. There is a nearly omni-directional pattern for the co-polarized component, which is desirable for Bluetooth applications.
[0055] It should be emphasized that the above-described embodiments of the present invention, particularly, any "preferred" embodiments, are merely possible examples of implementations, merely set forth for a clear understanding of the principles of the invention. Many variations and modifications may be made to the above-described embodiments of the invention without departing substantially from the spirit and principles of the invention. All such modifications and variations are intended to be included herein within the scope of this disclosure and the present invention and protected by the following claims.